Modem instructions sequentially alternating executions between sending a predetermined number of symbols by a transmit sequencer and receiving by a receive sequencer

ABSTRACT

A modem is implemented using a host processor or DSP and has execution code for implementing respective receive and transmit functionality which execute without any branching. Coordination of receive and transmit functionality is achieved using state indications in a common data area. Operation without branching eliminates the need for machine cycles associated with branching.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is related to U.S. patent application Ser. No.08/832,622 filed Mar. 31, 1997, by inventors Jim Bader, Scott Deans, RobMiller, Richard P. Tarquini, Bankim Wani and Jack Waters, entitled“CONTROLLERLESS MODEM”.

This application is related to U.S. patent application Ser. No.08/775,769 filed Dec. 31, 1996, by inventor Guozhu Long, entitled“PRECODING COEFFICIENT TRAINING IN A V.34 MODEM”.

It is also related to:

U.S. patent application Ser. No. 09/160,332, still pending filed Sep.25, 1998, by inventors Amir Hindie and Karl Leinfelder, and entitled“MODEM USING A DIGITAL SIGNAL PROCESSOR AND A SIGNAL BASED COMMAND SET.”

U.S. patent application Ser. No. 09/160,578, still pending filed Sep.25, 1998, by inventors Amir Hindie and Karl Leinfelder, and entitled“MODEM USING A DIGITAL SIGNAL PROCESSOR AND SEPARATE TRANSMIT ANDRECEIVE SEQUENCERS.”

U.S. patent application Ser. No. 09/160,571, still pending filed Sep.25, 1998, by inventors Amir Hindie and Karl Leinfelder, and entitled “AMODEM USING BATCH PROCESSING OF SIGNAL SAMPLES.”

U.S. patent application Ser. No. 09/160,570, still pending filed Sep.25, 1998, still pending by inventors Amir Hindie and Karl Leinfelder,and entitled “A MODEM WITH CODE EXECUTION ADAPTED TO SYMBOL RATE.”

U.S. patent application Ser. No. 09/160,569, still pending filed Sep.25, 1998, by inventors Wesley Smith, Karl Nordling, Amir Hindie, KarlLeinfelder, Sebastian Gracias and Jim Beaney, and entitled “INTEGRATEDAUDIO AND MODEM DEVICE.”

U.S. patent application Ser. No. 09/160,331, still pending filed Sep.25, 1998, by inventors Sebastian Gracias and Jim Beaney, and entitled“CODE SWAPPING TECHNIQUES FOR A MODEM IMPLEMENTED ON A DIGITAL SIGNALPROCESSOR.”

U.S. patent application Ser. No. 09/160,572, still pending filed Sep.25, 1998, by inventors David Pearce, Wesley Smith, Karl Nordling, AmirHindie, Karl Leinfelder, Sebastian Gracias and Jim Beaney, and entitled“A MULTI-MODEM IMPLEMENTATION WITH HOST BASED AND DIGITAL SIGNALPROCESSOR BASED MODEMS.”

U.S. patent application Ser. No. 09/160,587, still pending filed Sep.25, 1998, by inventors Guozhu Long and Jim Beaney, and entitled“SYNCHRONIZATION TECHNIQUES USING AN INTERPOLATION FILTER.”

U.S. patent application Ser. No. 09/160,577, still pending filed Sep.25, 1998, by inventors Guozhu Long and Jim Beaney, and entitled “A MODEMWITH A FAST GAIN TRACKER.”

U.S. patent application Ser. No. 09/160,538, still pending filed Sep.25, 1998, by inventor Jim Beaney, and entitled “A TONE DETECTOR FOR USEIN A MODEM.”

BACKGROUND OF THE INVENTION

1. Field of the Invention

The invention relates to communication modems and more particularly tomodems implemented using a Digital Signal Processor having simplifiedexecution code.

2. Description of Related Art

The use of modems to transmit digital signals across an analog channel,such as a telephone line, is well known in the art. Modem capabilitiesand performance have increased dramatically as the digital technologyutilized to handle information has exploded with a variety of newapplications and with large quantities of content. This technologyexplosion has resulted in increasing complexity for modems required tohandle increasingly complex protocols.

Recent generations of modems utilize different signaling rates atdifferent times or stages during their operation. Typically, a modemutilizes a dedicated processor or controller to carry out the operationsrequired for modem transmission and reception. Software which drivessuch dedicate processors is often convoluted, containing many branchesand jumps. Frequently, the dedicated modem processor is controlled by asequencer implemented as a finite state machine. The state of the finitestate machine changes as samples arrive and are sent in such a way as toimplement the modem functionality. Typically, incoming signals from ananalog channel are sampled by an analog to digital coder/decoder (codec)and signal samples are processed as they arrive from the codec. Thisimposes certain demanding real time performance requirements sinceprocessing of a given sample must be completed by the time the nextsample arrives.

Incoming signal levels to a modem are often adjusted by an automaticgain control (AGC) circuit. However, incoming signals are often subjectto a line “hit” which causes a momentary deviation from the desired gainlevel which cannot be compensated for by the AGC circuit.

Controllerless modems are also known which run as a separate process onthe host which they service. An example of such a controllerless modemis shown in the referenced co-pending application.

Modern computers are processing real time audio in digital form more andmore frequently. This audio processing can take the form of, forexample, telephone applications, stored audio files, audio filesaccompanying real time motion images and the like. Often, thisprocessing is ongoing at the same time as modem functions are occurring.

Digital signal processors are also known. These are relatively memorylimited devices which are designed for high performance processing ofdigital signals. They typically operate as an adjunct to the hostprocessor and can be configured to receive and handle processingassignments from the host computer and then return the results either tothe host or to a memory location specified by the host. Digital signalprocessors are now available which handle multiple streams of digitalsignals.

A variety of techniques are used to adjust the timing of two digitalsignal streams so that important events from both streams coincide intime. These techniques are called synchronization techniques.Frequently, synchronization is required when undertaking modemapplications or digital signal processing applications.

BRIEF SUMMARY OF THE INVENTION

The invention is directed to modem technology and to improvements thatare achieved by implementing modem code so that it executes withoutbranching.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 a block diagram showing an architecture suitable for integratingaudio and modem functionality in accordance with one aspect of theinvention.

FIG. 2 is a block diagram illustrating swapping of code between a hostand a DSP in accordance with one aspect of the invention.

FIG. 3 is a flow chart of a process for swapping code in and out of aDSP in accordance with one aspect of the invention.

FIG. 4 is a high level block diagram of an exemplary modem carrying outaspects of the invention.

FIG. 5 is a block diagram of an exemplary data encoder illustrated inFIG. 4.

FIG. 6 is a block diagram of an exemplary training encoder illustratedin FIG. 4.

FIG. 7 is a block diagram of an exemplary transmit engine illustrated inFIG. 4.

FIG. 8 is a block diagram of an exemplary receive engine illustrated inFIG. 4.

FIG. 9 is a block diagram of a far-near (F-N) echo canceller shown FIG.8.

FIG. 10 is a block diagram of a receiver shown in FIG. 8.

FIG. 11 is a block diagram of an equalizer having a fast gain trackershown in FIG. 8.

FIG. 12 is a block diagram of a listener (L-echo) echo canceller shownFIG. 8.

FIG. 13 is a block diagram of a debwarp and noise whitening filter shownin FIG. 8.

FIG. 14 is a block diagram of a phase corrector shown in FIG. 8.

FIG. 15 s lock diagram of a sync recovery circuit shown in FIG. 8.

FIG. 16 is a block diagram of a tone detector shown in FIG. 8.

FIG. 17 is a block diagram of a data decoder illustrated in FIG. 4.

FIG. 18 is a block diagram of a training decoder shown in FIG. 4.

FIG. 19 is a block diagram of an alternative phase corrector to thatshown in FIG. 14.

FIG. 20 is a diagram used to illustrate the operation of carriertracking as it occurs in FIG. 19.

FIG. 21 is a diagram of yet another alternative phase corrector shownFIGS. 14 and 19.

FIG. 22 is an illustration of an exemplary sample command format used inaccordance with one aspect of the invention.

FIG. 23 is a list of commands used in carrying out one aspect of theinvention.

FIG. 24 is a block diagram of the memory space of a DSP in accordancewith one aspect of the invention.

FIG. 25 is a flow chart of a load and execution sequence of modules on aDSP in accordance with one aspect of the invention.

FIG. 26 is a flow chart showing more detail of the transmit init moduleshown in FIG. 25.

FIG. 27 is a flow chart showing in more detail the receive unit moduleshown in FIG. 25.

FIG. 28 is a flow chart showing more detail of a Transmit Sequencershown in FIG. 25.

FIG. 29 is a flow chart showing more detail of a Receive Sequencer shownin FIG. 25.

FIG. 30 is a block diagram of an alternative memory space arrangement tothat shown in FIG. 24.

FIG. 31 is flow chart of a process for batch processing of receivedsamples in accordance with one aspect of the invention.

FIG. 32 is a flow chart of a process for batch processing of samples tobe transmitted in accordance with one aspect of the invention.

FIG. 33 is a flow chart of a process for controlling modem processingfunctions based on received symbol rate.

FIG. 34 is a block diagram showing use of both host based controllerlessmodems and DSP based modems.

DESCRIPTION OF THE INVENTION

FIG. 1 is a block diagram showing an architecture suitable forintegrating audio and modem functionality in accordance with one aspectof the invention. This diagram is symbolic in that it shows a separatehost domain 100 separated from a bus device domain 110 by a bus, such asPCI bus 105. A plurality of devices on the bus can be managed by thehost. Typically, each of those devices will have a device driver 120which serves as the interface to that device. The interface of each ofthe virtual device drivers 120 to the PCI bus 105 is through a streamprocessing virtual device driver 130 which can manage the plurality ofstreams originating and terminating in virtual device drivers 120. Anumber of devices may be connected to the PCI bus 105. One such device,namely device 140, is shown in FIG. 1. Device 140 is a digital signalprocessor capable of handling multiple digital streams. Preferably, themultiple stream digital signal processor is a DSP identified as CS4610available from Cirrus Logic.

The digital signal processor has a stream processing operating system150 which manages a variety of tasks which can be run either inforeground, midground or background (160). The digital signal processormanages a plurality of serial ports 170 and a plurality of generalpurpose input/output ports 175. Connected to the serial ports 170 is anaudio codec array 180. The audio codec array performs analog to digitalconversion and digital to analog conversion of analog signals from linedrivers 195 and from digital data arriving over a serial port 170,respectively. Many of the audio sources feeding audio codec array 180are high quality audio sources requiring an elevated sampling rate tomaintain the fidelity during the digital signal processing. A pluralityof codecs 185 form a modem codec array. Modem signals on a modem lineare sampled, converted to digital and applied to the digital signalprocessor over the serial ports 170. Digital information from thedigital signal processor can pass over serial port 170 to the modemcodec array which converts the digital into analog and applies theanalog output to the modem line. A telephone line connects to a dataaccess arrangement (DAA). The DAA serves as a line interface between atelephone line and the DSP 140. The signalling portion of telephonesignals is handled in the DAA and control lines between the DAA and thegeneral purpose input/outputs 175 are used to passing ringinginformation to the DSP and receive control signals from the DSP. Thenon-signaling portion of the information is applied to the modem codecarray as another modem input signal, where it is sampled and applied tothe serial ports 170 like any other modem signal.

FIG. 2 is a block diagram illustrating swapping of code between a hostand a DSP in accordance with one aspect of the invention. A host memoryspace 200 is shown and a corresponding memory space 210 for the DSP isalso shown. A DMA controller 220 can manage a transfer of informationfrom the host memory space 200 to and from the DSP memory space 210without host intervention. The DMA controller can also be arranged totransfer information from mass storage 225 into the host memory spaceand back. In some digital signal processors, where memory space islimited overall or in which memory space is limited for use by aparticular application, it may be desirable to load and execute only themodules of code that are necessary for a particular state of the modem.A plurality of such modules 230 is stored in host memory space 200. Asdiscussed more hereinafter, when the DSP is initialized with a modemapplication, a pre-init module is loaded into the DSP and stays residentuntil the modem application completes. The pre-init is accompanied by atransmit sequencer and a receive sequencer which also stay resident.Associated with each modem application is a swappable space in the DSPwhich can be utilized to bring in and execute one or more of the modules230 as may be required for modem execution. Once a particular modulefinishes executing, it can be overwritten by a DMA transfer of anothermodule to be executed in the DSP. FIG. 2 illustrates only the transferof code modules for a modem application in and out of the DSP memoryspace. This figure does not illustrate signal processing, but only theuse of code swapping to implement the modem applications in the DSP.Signal processing is discussed more hereinafter.

FIG. 3 is a flow chart of a process for swapping code in and out of aDSP in accordance with one aspect of the invention. When a modemapplication is required, the base code including a pre-initializationsequence is loaded to the DSP memory space (300). The Rxinit and Txinitmodules are swapped in from the host memory space into the DSP memoryspace and are executed (310, 320). The Rxinit and Txinit processestablished the transmit and receive sequencers in the DSP memory spaceand prepares them for handling modem functions. The encoder-init anddecoder-init modules (330, 340) are swapped in, run and swapped out.These set up the initial operating conditions for the various functionalblocks of the modem, discussed more hereinafter. Once the initializationphase is completed (after 340), encoder run and decoder run processesexecute sequentially and continuously to process the signal samplesgoing to and from the codecs.

The encoder run and decoder run modules are each designed to do batchprocessing on a group of symbols. Typically, the symbols will beprocessed in eight symbol batches. This reduces the processingrequirements considerably over signal sample oriented processing. Aplurality of signal samples is normally required in order to identifyeach symbol.

Functional Description

FIG. 4 is a high level block diagram of an exemplary modem carrying outaspects of the invention.

The modem operates in two modes, a training phase and a data phase.During the data phase, bitstream data from the host is encoded intosymbols by the data encoder (400). The symbols are then modulated andfiltered into samples by the transmit engine (410). These samples aretransferred back to the host (to be transmitted on the line). The hostmanages the sample rate conversion and the code. On the receive side,samples from the host (received from the line) are de-modulated andfiltered into symbols by the receive engine (420). The receive engineuses symbols from the transmit engine to perform the echo-cancellationand uses symbols from the decoder to do channel equalization and samplesfrom the receive base band filter to do listener echo cancellation. Thesymbols are decoded by the data decoder (430) into bitstream data whichis transferred back to the host.

During a training phase, the modem sequencer triggers the trainingencoder (440) to generate the appropriate training sequences. Thesymbols generated by the training encoder go through the same transmitengine as in the data phase. Similarly, the training decoder (450) isfed symbols by the receive engine. The sequencer also controls variousparameter settings in the transmit and receive engine, like step sizes,etc. which are different in the training and data phases. Initially, thesequencer connects the transmit/receive engines to the trainingencoder/decoder. After training is complete, the engines are switched tothe data encoder/decoder. The sequencer re-enters the training phase, ifrequested by the host controller.

Data Encoder

FIG. 5 is a block diagram of an exemplary data encoder illustrated inFIG. 4. Bitstream data from the host is scrambled (500) to randomize it.The data framer (510) distributes the incoming bits 1 into separatestreams to the shell mapper (520), the constellation mapper (530), thedifferential encoder (540) and the subset labeler (550). The shellmapper maps the input bits into 8 ring indices. These indices are usedto pick the rings in the constellation used by the next 8 symbols. Theconstellation mapper uses the ring index and bits from the data framerto pick the appropriate point in the constellation. This point is thenrotated by 0, 90, 180 or 270 degrees (555), depending on the input fromthe subset labeler. The subset labeler uses information from thedifferential encoder and the trellis encoder (560) to pick the desiredrotation angle. The symbol is pre-coded (570) to aid the remoteequalizer and subjected to a non-linear transfer function (to combatnon-linear distortion on the channel). The symbol is used by thetransmit engine to generate the samples to be transmitted.

Training Encoder

FIG. 6 is a block diagram of an exemplary training encoder illustratedin FIG. 4. The training encoder encodes the various training sequencessent by the modem in phase 3 and phase 4. The timing and order ofgeneration of the training sequences is controlled by the overall modemsequencer. A brief description of the various sequences generated is asfollows.

PP sequence: This is a constant amplitude zero auto-correlation sequencesent for fast training of the equalizer. It consists of six periods of48 symbols and is sent in training phase 3.

S/S sequence: This consists of two alternating points in the four pointconstellation and a phase reversed version. It is used as a marker inboth phase 3 and 4.

TRN sequence: This consists of a sequence of ones. It is used fortraining in both phase 3 and 4.

J sequence: This is a 16 bit pattern which specifies whether phase 4training will use a 4 point constellation or a 16 point constellation.

J′ sequence: This is a 16 bit pattern and is used to indicate thebeginning of phase 4.

E sequence: This is a 20 bit sequence of ones. It is used to indicatethe end of phase 4.

MP sequence: This sequence is an 88/188 bit pattern with a 16 bit headerand 15 bit CRC. It is used by the modems to exchange data phaseparameters like bit rate, precoder coefficients, etc. All sequences,except PP and S/S are scrambled and differentially encoded before beingmapped to symbols. The sequence B1 which is sent just before data phaseis part of the data phase initialization.

Transmit Engine

FIG. 7 is a block diagram of an exemplary transmit engine illustrated inFIG. 4. Symbols from the data encoder are up-sampled to 3×the symbolrate. The base-band filter (BBF 700) also serves as the up-samplinganti-aliasing filter. The BBF is a 48-tap, real FIR filter. The filteredsamples are then passed through a pre-emphasis filter (710) which doesthe pass-band spectral shaping specified by the remote modem in thetraining phase. There are 11 sets of 6-tap complex FIR filters, one foreach of the specified spectral shapes. The samples are then modulated bythe carrier (720) before being transferred by the host for finalup-sampling and transmission.

Receive Engine

FIG. 8 is a block diagram of an exemplary receive engine illustrated inFIG. 4. Input samples from the host (received on the channel) are firstpassed through the F-Necho canceller (800). The cancellers remove thenear and far end echo of the transmitted signal from the receivedsignal. The signal is then de-modulated and filtered by the receiver(810). The receiver uses the Tx−Rx time difference computed by the syncrecovery section (820) to match the receive sample time to the transmitsample time of the remote modem. The distortion introduced by thechannel is removed by the equalizer (830). The equalizer uses errorsymbols from the phase corrector (840) for tap-update and the rotatederror symbols for gain tracking.

The listener echo, which is the echo of the received signal itself isremoved (850) before the signal is passed through the non-linear decoderand noise-whitening filter (860). The phase corrector tracks the phasejitter and the frequency offset of the carrier and corrects for phaseerrors introduced. The corrected symbols are then sent to the datadecoder or the training decoder for final decoding to bitstream. Duringdata phase, a tone detector (870) is used to detect requests from theremote modem for rate negotiation or retrain.

F-Necho Canceller

FIG. 9 is a block diagram of a far-near (F-N) echo canceller shown inFIG. 8. The echo cancellers remove near-end and far-end echos of thetransmitted signal from the received signal. The received samples fromthe host are passed through a hum filter (905) which removes DC andpower-line hum. The near-end and far-end echo is then subtracted fromthe signal. This subtraction is performed on double precision samples.The samples are then sent to the receiver for de-modulation andfiltering. The echo is generated using symbols from the transmit engine.

Symbols from the transmit engine are pre-modulated (900) to thepass-band and fed to the bulk delay line (910) and the Necho filter(920). The bulk delay line matches the delay experienced by the far endecho on the channel. The Necho is generated by passing the symbolsthrough the 2 120-tap real adaptive FIR filters operating on the realand complex parts of the symbol respectively. Since the received samplesare at 3×symbol rate, 40 taps of the canceller are used and updated ateach sample. The Fecho is generated in a similar fashion using anotherset of 120 tap real adaptive FIR filters (930).

A Fecho carrier tracker (940) is used to correct for phase shiftsexperienced by the Fecho due to FDM (if any) on the line. The carriertracker measures the phase difference between the input and output ofthe Fecho filter and uses it to generate a phase correction (950)(in aPLL-type configuration).

Receiver

FIG. 10 is a block diagram of a receiver shown in FIG. 8. The receiverde-modulates the received samples to the base-band (still at 3×symbolrate) The AGC (1000) produces a near constant signal power by trackingthe gain of the channel. The gain is tracked by comparing the receivedsignal power to a fixed reference. The sampling rate of the receivedsignal is locked to the local transmit clock and needs to be matched tothe remote transmit clock. The sample time adjust (1010) feeds two orfour samples to the de-modulator depending on whether the remote clockleads or lags the local transmit clock. The signal is then demodulatedinto the base band. The interpolator (1020) is a 3-tap FIR filter. Thefilter taps are set depending on the estimated difference between the Txand Rx clocks. The interpolated signal is band limited by the base bandfilter (1030). The receive base-band filter is a 48-tap FIR filter,identical to the transmit filter.

Equalizer

FIG. 11 is a block diagram of an equalizer, having a fast gain tracker,shown in FIG. 8. The distortion due to the channel is removed by theequalizer. The equalizer (1120) is a 60 tap complex adaptive FIR filter.The taps are updated (1110) twice every three samples using the errorsymbols from the phase corrector. The equalizer taps are used by thesync recovery section to estimate the Tx−Rx sample rate difference. Thegain tracker (1100) is used to track sudden changes in the gain of thechannel. The rotated error symbols from the phase corrector and thecurrent gain is used to update the new gain value.

Lecho Canceller

FIG. 12 is a block diagram of a listener (L-echo) echo canceller shownin FIG. 8. The Lecho canceller removes the echo of the received signalfrom the received signal. A 16-tap complex adaptive FIR filter is usedto cancel the listener echo (1200). The bulk delay line (1210) matchesthe delay experienced by the signal on the channel. The Lecho cancellertaps are updated using the error signal from the phase corrector.

Dewarp, NWF

FIG. 13 is a block diagram of a dewarp and noise whitening filter shownin FIG. 8. The dewarping and noise whitening filters do the inverseoperation of the non-linear encoder and filter at the remotetransmitter. The non-linear decoder (1300) uses a polynomial of degree 4as an approximation to the inverse of the non-linear encoding functionspecified in the standard. It also scales the signal to the slicinggrid. The noise whitening filter (1310) is a 3 tap complex FIR filter,whose co-efficients are sent to the remote modem during the trainingphase. The co-efficients are trained using a complex version of theLevinson-Durbin algorithm. An example of this is shown in co-pendingU.S. application Ser. No. 08/775,769 referenced above.

Phase Corrector

FIG. 14 is a block diagram of a phase corrector shown in FIG. 8. Thephase corrector corrects for the frequency offset in the carrier and thecarrier phase jitter. The carrier frequency offset is tracked by asecond order PLL (1400), which uses the phase error (1410) between thesymbols from the decoder and the input symbols. The phase jitter istracked by a 60 tap real adaptive FIR filter (1420). Since the jitter isnot very large, ⅓^(rd) of the filter taps are updated each symbol.

The amplitude error between the output of the phase corrector and thedecision symbols from the decoder is sent to the gain tracker in theequalizer. The decision symbol from the decoder is then rotated (1420)by the same amount as the input symbol, but in the reverse direction.The error between the rotated decision symbols and the input symbols isused to update the equalizer taps.

Sync Recovery

FIG. 15 is a block diagram of a sync recovery circuit shown in FIG. 8.The sync recovery section estimates the frequency difference between thelocal and remote sample clocks. This delay will appear as a constantfrequency offset introduced by the channel and reflected in theequalizer taps. To compute this constant shift, a DFT (1500) is done onthe equalizer taps to generate the B/4 and −B/4 spectral components,where B is the baud rate. The phase difference between the components isthe sync error, which is filtered through a second order PLL (1510) toget the time difference between the clocks.

Tone Detector

FIG. 16 is a block diagram of a tone detector shown in FIG. 8. Theremote modem sends tone A (answer modem) or B (call modem) to initiate aretrain. The modem is supposed to go to training phase 2 on receipt ofthis tone. The tones are at the frequencies 2400 Hz and 1200 Hzrespectively. Since the normalized frequency will be different for thevarious symbol rates, an adaptive tone detector (1600) is used to detectthese tones. The detector is a complex adaptive FIR filter of the form1-z⁻¹. If a tone is being transmitted the tap will converge to thefrequency of the tone. The amplitude of the tap is used to detectwhether the tone is being transmitted.

The remote modem sends the sequence S to initiate a rate negotiation.The modem is supposed to go to training phase 3 on receipt of thissequence. The spectrum of the sequence S has peaks at the three complexroots of 1. It is detected by comparing the energy at the input andoutput of a notch filter which has zeros at these frequencies. The notchfilter used is a simple FIR filter of the form 1-z⁻⁶.

Data Decoder

FIG. 17 is a block diagram of a data decoder illustrated in FIG. 4. Thedata decoder section converts the symbols into bitstream data. Itperforms the inverse of the operations done by the data encoder. TheViterbi decoder (1700) only supports the 16 state convolution code atthe remote encoder. The Viterbi decoder picks the path through thetrellis with the maximum likelihood. The final decision is generatedafter a delay of 16 4D symbols, i.e. 32 symbols. A zero-delay 4Ddecision, i.e. a delay of 1 symbol is used for the equalizer tap update.After preceding reconstruction (1710), the ring indices, the uncodedbits and the differentially encoded bits are extracted and packedappropriately before transferring them to the host.

Training Decoder

FIG. 18 is a block diagram of a training decoder shown in FIG. 4. Thetraining decoder decodes the received symbols into received sequences.These sequences are then compared against the desired received sequencesin order to trigger error procedures.

FIG. 19 is a block diagram of an alternative phase corrector to thatshown in FIG. 14 that will be used in the following discussion.

V.34 Phase & Amplitude Correction

1. Introduction

As an example, this section will discuss the implementation of carriertracker, phase jitter tracker, and amplitude jitter tracker for V.34.Although the names of the submodules are similar to their V.32biscounterparts, due to a very large constellation and high performancerequirement for V.34, the actual implementation has to be modified andimproved significantly, as discussed herein.

At the equalizer output, there is an adaptive gain tracker (1900) tobring signal towards slicing grid. This is especially useful for V.34since there are many possible signal constellations and preceding mayalso change the signal power. As a result, the received data signalpower may not be exactly equal to the power in training. Regular AGC isnot fast enough to adjust. After the gain adjustment, the signal passesthrough dewarper (1910--nonlinear decoder) and noise whitening filter,and is scaled to 80H grid. ROTOR (1920) rotates the signal to removecarrier frequency offset. JTOTOR (1930) rotates the signal again toremove phase jitter. Then the signal is sent to Viterbi decoder (1940).The Viterbi decoder in V.34 is 4D-based, and the final decision isdelayed by 16 4D intervals (32 bauds). If we compute the error signalsbased on the final decision, the errors will have a long delay, and wehave to use the delayed least mean square LMS algorithm to update allthe adaptation loops. This is possible, but not convenient. A differentapproach is used here. In addition to the regular Viterbi decoderdecision, we compute an early decision—a delay 0 decision. Namely, inthe Viterbi decoder, before we trace back, we make a decision for thecurrent 4D, i.e., at the end of each 4D, we have the early decision forthe two 2D symbols in this 4D. Since the decision for the first 2D inthe 4D is not available until the second 2D, we have one-baud decisiondelay (delay 1 decision in the figure). Therefore, we still have to useDLMS, but its delay is only one baud. In the figure, each “D” blockmeans one baud delay, and they are necessary in the DLMS algorithm.

Next, each function unit will be discussed in detail.

2. Carrier Tracker (FIG. 20)

Note that the constant β_(—)2 should be inversely proportional to thebaud rate. It is initialized in the beginning of phase 3. In the V.34implementation, both f and τ have double precision to improve theaccuracy. In V.32bis code, only f uses double precision.

A new algorithm for computing the phase difference between two complexvectors is implemented for v.34. This algorithm is different from theone used in V.32bis, where the phase difference α between two complexvector S and D is computed as follows.

α≈sin α=(S _(r) *D _(i) −S _(i) *D _(r))/¦D¦¦S|≈(S _(r) *D _(i) −S _(i)*D _(r))/¦D¦ ²

where α is the angle between D and S. A look-up table for 1/¦D¦ isprestored in memory to facilitate the computation. This is feasible inV.32bis since the signal constellation is limited. As a result, such atable is quite small. This is no longer feasible for V.34. V.34 has asignal constellation up to 1664. Furthermore, the preceding may expandthe constellation even further. Even though we store only a quarter ofit, we still need a large memory space, and its dynamic range is big.

A new algorithm is as follows.

x=tan α=(S _(r) *D _(i) −S ₁ *D _(r))/(S _(r) *D _(r) +S _(r) *D _(r))

α=arctan(x)≈0.999960426x−0.331837378x**3+0.184496733x**5−0.079428803x**7

Note that here we have to make sure that −1 ≦x′<1, i.e., −45°≦α<45°0.Hence, we can scale a such that 45° becomes 7ff. Note that this scale is4 times bigger than the normal scaling where 7fff means 180°. Thislarger scale is very helpful in improving jitter tracker and carriertracker precision. This new algorithm is more accurate than the one inV.32bis. The subroutine for this algorithm is called several times tocompute the phase errors between different signals. After we get theangle, we can call another subroutine to compute the sine and cosinevalues, which will be used in the rotation. This subroutine is alsobased on power series expansion, hence no prestored table is needed.Note that the scaling for the phase error is 4 times bigger, henceshould be shift right by 2 bits before calling sin/cos routine.

3. Phase Jitter Tracker:

Phase jitter is compensated in JROTOR, which rotates the signal by anangle θ, which is the output of an adaptive FIR filter (phase jitterestimator), whose input is jeph, which is the phase error between thedelayed ROTOR output and delay 1 decision. The phase error jeph isactually the same as eph for the 2nd order PLL discussed above. Thecorrection angle θ can be considered as the linear prediction of thephase error for the current baud signal based on the previous phaseerrors. Note that we compute the phase jitter estimator output θ beforewe shift in new phase error jeph.

To update the phase jitter estimator coefficients, we compute the phaseerror between the delayed JROTOR output and delay 1 decision. The phaseerror obtained is named jitt_err, and is used to update the tapcoefficients.

Since phase jitters are usually very small, jout is typically verysmall. Also, jit_out is even smaller.

Here it is very important to fully utilize the available dynamic range.An effective technique is to scale down the filter output before storingto jout. In that way, the filter output should be bigger, so are thefilter tap coefficients. Thus, for the same adaptation noise in jout, wecan use a bigger step size so that underflow can be avoided. In theprevious V.32bis implementation, the filter output is shift-left twicebefore storing to jout. This makes the situation even worse. In the newimplementation, the filter output is shift-right twice, and the accuracyhas been improved considerably.

In V.32bis code, the adaptive phase jitter estimator has 60 taps. Sincethe phase jitter frequency is no larger than 300 Hz, it is possible toreduce the number of taps by down sampling. In V.34 implementation, wedown sample the input jeph by 3, thus reduce the number of taps to 20for the same filter span. The input delay line is still the same (64long) since we have to compute the output once every baud.

For the tap coefficient updates, since there is one baud decision delay,we have to use the DLMS algorithm. This is implemented by moving backthe input buffer read pointer by 1 step in the adaptation.

4. Adaptive Gain Tracker

In the figure above, there is an adaptive gain tracker to adaptivelyadjust the equalizer output gain when there is a sudden signal levelhit. Besides the normal gain hit from the channel, it is very useful inthe beginning of B1 of the V.34 standard. Normally, the AGC beforeequalizer normalizes the signal power. However, the signals are rescaledin nonlinear decoder and after noise whitening filter. Modems fromdifferent manufacturers may differ slightly in the scaling. Althoughequalizer coefficient update can bring signal after scaling to theslicing grid, such an adaptive gain tracker is much quicker to adjust.Note that B1 lasts only 35 or 40 ms, which is typically not long enoughfor normal equalizer adaptation.

The gain tracker (FIG. 11) is turned on in the beginning of B1. If thereceived signal power or the scale value in the receiver is slightlyoff, the signal point at the input to the Viterbi decoder will be offfrom the constellation grid. Since the phase errors are alreadycompensated, the signal point and the decision point should ideallydiffer by the magnitude plus noise. We compute the approximate magnitudeerror as follows:

e(n)=[abs(D _(r)(n))−abs(S _(r)(n))]+[abs(D _(i)(n))−abs(S _(i)(n))]

where D(n) and S(n) are the decision point and the signal point,respectively, and the subscript r and i mean the real and imaginarypart, respectively.

Note that when the gain is too small, e(n) should be positive while ifthe gain is too big, e(n) should be negative. We then use e(n) to updatethe gain g(n):

g(n)=(1−w)*g(n−1)+c*e(n)

where w is a leakage constant and c is the step size. g(n) is used toscale the equalizer output y(n):

y′(n)=y(n)(1+g(n))

g(n) is initialized to 0. If there is a sudden gain hit, g(n) willchange quickly to its proper value so that mean square value of e(n) isminimized. A small leakage constant w is used to enhance the numericalstability. After the gain is stabilized, g(n) will slowly leak to zeroor a small fixed value. The gain hit is transferred gradually to theequalizer coefficients.

If AGC is running at the same time, it will adjust the signal powerslowly to the nominal value, and g(n) will track the signal poweraccordingly.

5. Amplitude Jitter Tracker

The adaptive gain tracker can compensate gain hit properly. However, itis not designed for compensating the amplitude jitter. In V32bis, thereis an amplitude jitter canceller. It is not implemented for V.34 now.However, a design is provided below.

Firstly, the amplitude error is determined.

In order to update the adaptive phase jitter estimator, one computed thephase error between the delayed JROTOR output and delay 1 decision.Based on this phase angle, one can rotate the delayed JROTOR outputsignal exactly towards the delay 1 decision. After the rotation, the twosignals have exactly the same phase, and they differ only by themagnitude. We can simply subtract one from the other to obtain theamplitude error vector.

Note that the amplitude error vector is a complex vector. We can computeits magnitude as the amplitude error. We can simply compute the sum ofthe absolute values of its real and imaginary parts as the amplitudeerror. If necessary, more accurate approximation of the magnitude can beobtained by:

¦E¦max(E _(r) ,E _(i)))+0.3.006 min(E _(r) ,E _(i))

In V32bis implementation, the error signal amperr then passes through alow-pass filter, whose output is used to scale the signal before ROTORto compensate the amplitude jitter. The low-pass filter is as follows:

amp_int=amp_int+(0.5-amp_int)/128+amperr*{fraction (3/16)}.

It is a one-pole IIR integrator with time constant about 128 bauds.

The performance of the amplitude jitter tracker has been tested. Whenthe amplitude jitter canceller has to track a low-frequency sinusoidamplitude jitter, the performance is not good enough to compensate theamplitude jitter, because the low-pass filter can only filter out thelow frequency jitter, but can not adjust the output phase to exactlymatch the low frequency jitter in signal. A better design is to add anadaptive amplitude jitter estimator similar to the phase jitterestimator. Namely, we can use the previous amplitude errors to estimatethe amplitude error for the current baud. Such a structure is shown inthe figure below. Note the amplitude jitter is compensated at the outputof equalizer so that we don't need an inverse for the equalizer errorcalculation. This makes the implementation more efficient. Such anadaptive amplitude jitter canceller needs some memory space and cycles,and may not be desired.

6. Error Signal for EO update

In FIGS. 19 and 21, one has shown two different alternatives forcalculating error signal for updating equalizer. In the first figure,the error signal is calculated at the output of noise whiterning filter,while in the second figure, the error is computed at the equalizeroutput. The 2nd approach seems to be ideal, however, this approach mayhave error propagation problem in the inverse noise whiterning filtersince it has an IIR structure. Therefore, in actual V.34 implementation,the first approach has been used.

FIG. 22 is an illustration of an exemplary sample command format used inaccordance with one aspect of the

We also have to find out the sign of the amplitude error. Note that theamplitude error vector and the decisions vector are always on the sameline, i.e. the angle between these two vectors is either 0 degrees or180 degrees. If we compute the dot product of them, the result will bepositive if the angle is 0 degrees and negative if the angle is 180degrees. We then combine the amplitude error calculated above with thissign information to form the final amplitude error. Suppose one denotesthe delayed JROTOR output signal by S, the delay 1 decision by D and theamplitude error vector by E we can write:

S′=Sd ^(j(jit) ^(_(—)) ^(err)) E=D−S″amp_err=sign[Er, *Dr+Ei*Di]*(¦E_(r)¦+¦E¦)

where the subscript r and I denote the real and imaginary part,respectively.

Note that this algorithm is completely different from the one used inV.32bis, where the amplitude error is computed as follows.

E=D−S

amperr=¦D¦−¦S¦≈¦D¦−¦S¦cosα=(¦D¦ ² −¦D¦¦S¦cosα)/¦D|=(D_(r) *D _(r) +D_(i) *D _(i) −S _(r) *D _(r) −S _(i) *D _(i))/¦D¦=(E _(r) *D _(r) +E_(i) *D _(i))/¦D¦

where α is the angle between D and S.

Since the look-up table 1/¦D¦ is no longer available in V.34 (it is toobig to store), it is hard to use that formula. Note that there is anapproximation ¦D¦−¦S¦≈¦D¦−¦S¦ cos α in that implementation. As a result,if there is a phase jitter, and no amplitude jitter, such anapproximation will show a false amplitude jitter. In the new algorithm,there is no such an approximation, and one does not need the table for1¦D¦.

The simplified amplitude error calculation algorithm discussed in theadaptive gain tracker may also be used. It is much simpler, and quiteeffective. invention. The preferred command format includes a mnemonic2200 together with a signal table index 2210 and one or more parameters2220. The particular mnemonics utilized in generating code for the modemapplication modules are part of a language that is customized for modemapplications. That is, the mnemonics cover commands relating to thevarious types of signals that need to be generated or processed ratherthan utilizing general programming commands. An op-code is associatedwith each mnemonic and the modem functions programmed using the modemspecific language as pseudo code can be either interpreted or compiledto run as machine code on the DSP.

A list of high level state machine commands useful in implementing amodem or signal based language is shown in FIG. 23. These commandsprovide an extremely powerful language which expresses the generationand recognition of the necessary signals for carrying out a modemprotocol. This greatly facilitates development time of modemapplications and reduces code size through a terseness of expressionthat is very powerful.

FIG. 24 is a block diagram of the memory space of a DSP in accordancewith one aspect of the invention. The DSP memory space 2400 includes alibrary of commands 2410 which specify the actions to be taken by themodule when a particular command is asserted. There is a data area 2420which contains a data structure for each of the modules. The datastructure has two components. The first is a parameter portion MPB andthe second is a data result portion MDB. As each module is called, thesequencer extracts needed information from the data area and passes anMPB data structure containing the parameters required for the call tothe module. The module executes and returns an MDB portion of the datastructure containing the results of execution. Thus, all data requiredfor execution of the modem functionality is contained in a separate dataarea and is selectively extracted for use in execution of the modules.All state information is contained within the data area. Thus, as eventsoccur in the DSP, the contents of particular fields of the data area maychange, which then results in changed data when a module is called forexecution.

A particularly advantageous way of arranging the programming flow forthe DSP when using a modem application involves the way in which thetransmit sequencer and receive sequencer are generated. Each of thesemodules is implemented without any branching. That is, every statementis executed every time in the same sequence. As shown in FIG. 24, asreflected in FIG. 3, the transmit sequencer is executed and thenfollowed by the receive sequencer on a repetitive ongoing basis. Thus,the transmit sequencer will execute every one of its statements and thenthe receive sequencer will execute every one of its statements and thenreturn to the transmit sequencer for execution of each one of itsstatements again and so on. The combination of batch processing ofsymbols, the avoidance of any branching or conditional commands togetherwith the arrangement of all data in a common data area in memory ensuresvery high performance execution of modem functionality.

FIG. 25 is a flow chart of a load and execution sequence of modules on aDSP in accordance with one aspect of the invention. When modem servicesare required on the DSP, it will be initiated by an operating systemcall (2500). When that occurs, the transmit init module 2510 and thereceive init module 2520 will execute setting up the respectivesequencers. Optionally at this point the transmit and receive initmodules have been overwritten or removed from the DSP providingadditional memory space. The transmit sequencer 2530 and the receivesequencer 2540 then operate substantially continuously in a loop untilthe state of the data in the data area indicates that the communicationshave ended. At which time they will terminate operation.

FIG. 26 is a flow chart showing more detail of the transmit init moduleshown in FIG. 25. When the transmit init module is called, it willinitialize the training encoder (2600), initialize the modulator (2610),initialize the echo canceller (2620) and initialize the analog transmitinterface (2630).

The functional modules discussed earlier in conjunction with theoperation of a modem are for the most part implemented in software. Thusthe initialization of these functional blocks is discussed in FIG. 26and subsequent figures involves the loading of the software for aparticular functional module and setting up the appropriaterelationships among the modules so that the modem functions can occur.Some of the modules interface with hardware devices. For example, thecodec arrays which provide the samples of signal levels on an incominganalog line and convert digital samples to analog for placement on thatline have a hardware aspect to them in that they are initialized andreadied for operation by software commands issued from, typically, adevice driver or equivalent.

FIG. 27 is a flow chart showing in more detail the receive unit moduleshown in FIG. 25. When the receive init module is called, it initializesthe train decoder (2700), the tone detector (2710), the receivedemodulator (2720), the equalizer (2730), the echo canceller (2740), thephase corrector (2750) and the analog receive interface (2760).

FIG. 28 is a flow chart showing in more detail the transmit sequencer(TxSequencer) shown in FIG. 25. The transmit sequencer calls the encoderfor processing one or more signal samples into symbols for transmission(2800). When that is done, the sequencer checks for a new state (2810)and then calls the modulator (2820) to prepare one or more symbols fortransmission (2820). The sequencer then calls the echo canceller toprovide the signal values necessary for echo cancellation in the modem(2830).

FIG. 29 is a flow chart showing in more detail the receive sequencer(RxSequencer) shown in FIG. 25. Like the transmit sequencer, the receivesequencer calls a plurality of modules in sequence. It first calls thereceive demodulator module. The receive demodulator will check to see ifthe number of signal samples received is equal to some number N. N can,of course, be 1 but preferably is larger than 1 to permit batchprocessing of the samples. Then the sequencer calls detect tone todetermine whether or not certain tones exist within the incoming signal.Typically, these are control tones such as S used in the V.34 sequences.The sequencer then calls the equalized (2910), recover signal (2915),cancel echo (2920), phase control (2925), parse new state (2930), phasecorrect update (2935), and disposition of data (2940) in sequence.

FIG. 30 is a block diagram of an alternative memory space arrangement tothat shown in FIG. 24. In this arrangement, transmit sequencer 2430 andthe receive sequencer 2440 do not operate sequentially but rather inparallel. Synchronization between the two running sequencers is achievedthrough reference to the data area. It is sometimes the case that, whenprocessing modem sequences such as V.34, that certain symbols or tonesmust be transmitted until a response is received from the receiver. Thusthe transmitter might be transmitting tone of a given duration which canbe interrupted when the proper tone or response is received from thereceiver. Since the DSP is a multi stream DSP, separate streams can beprocessed through the transmit sequencer and the receive sequencer sothat each can operate relatively independently of the other, except whenneeded.

FIG. 31 is flow chart of a process for batch processing of receivedsamples in accordance with one aspect of the invention. As discussedabove, the analog signal from the analog line is sampled utilizing acodec and the receive samples are stored in a buffer (3100). Once thenumber of samples received equal sum value N (3110) the receivesequencer will detect that the number of samples has reached the levelrequired for processing and will process those samples as describedabove. If the number of samples received is less then N, the receivesequencer will detect that and no samples will be processed. Althoughthis appears to be a branching operation, in implementation it is not.As the codec is serviced and samples are found in the buffer, the countof samples will be stored in the data area of the DSP and incremented asnew samples are received and processed. The receive sequencer will referto that value. If the value is less than N, subsequent steps will nothave arguments and nothing will occur. The next module will then run.However, if the number of samples received is N or more, there will beproper arguments for the code handling the received samples and thesamples will be processed as described above.

FIG. 32 is a flow chart of a process for batch processing of transmittedsamples in accordance with one aspect of the invention. This process isanalogous to that described in conjunction with FIG. 31.

FIG. 33 is a flow chart of a process for controlling modem processingfunctions based on received symbol rate. As discussed previously,various symbol rates are appropriate to corresponding phases or timeintervals of modem operation. For example, in various training phases,the symbol rate is specified to be a fixed value. However, whentransmitting data outside of the training phases, the symbol rate ispreferably much higher. This is one point where the symbol baselanguage, described previously, has considerable advantages. Bystructuring the modem functionality in terms of a symbol based language,when incoming samples are received from an analog line (3300) thosesamples will be converted to symbols (3310). Those symbols can arrivequickly or those symbols can arrive slowly depending on the symbol rateon the communications line. Since the modem functions are based onarriving symbols, the modem processing functions automatically adapt tothe symbol rate. If many symbols arrive in a given unit of time, theywill be processed quickly. If fewer symbols arrive in a given unit oftime, they will be processed more slowly. Thus, the signal processingoperations adapt automatically to the incoming symbol rate and executeonly when needed based on the arrival of the symbols. Thus, many of thedifficulties associated with programming modems to account for varioussymbol rates are eliminated. The modem processing functions adaptautomatically based on symbol rate because of the symbol processinglanguage utilized in the modem code.

FIG. 34 is a block diagram showing use of both host based controllerlessmodems and DSP based modems.

As discussed in the aforementioned co-pending application, one or morecontrollerless modems may be resident on the host and operate from thememory space of the host. Similarly, as described herein, one or moremodems may be loaded into the DSP and run from the memory space of theDSP. The fact that DMA transfer can be utilized to swap code in and outof the DSP as needed to operate modems suggest that not only individualmodules of modem code may be swapped in and out but in fact entire modemcode stacks could be swapped in and out of the DSP as required. Thiscreates a very flexible architecture in which a modem can be run eitherusing the host processor or using the DSP. If one desired to use acommon language for all modem code, differences in instruction sets forthe host and the DSP processors can be accommodated by different commandlibraries which would handle the common command language mapping to themachine instructions appropriate for the host or the DSP. In this way, avery flexible modem architecture is achieved in which a plurality ofcommunication functions can occur in an ongoing manner using theresources of either the host or the DSP or both.

There has thus been described a number of significant improvements inmodem technology which provide both high performance and excellentquality modem processing but ease of design and implementation as well.

Although the present invention has been described and illustrated indetail, it is clearly understood that the same is by way of illustrationand example only and is not to be taken by way of limitation, the spiritand scope of the present invention being limited only by the terms ofthe appended claims and their equivalents.

What is claimed is:
 1. A modem, comprising: a processor with a memorysystem which stores instructions for an encoder, a transmit engine, adecoder, and a receive engine wherein said instructions comprise:sending to a device to which the modem is to communicate and batchprocessing, by a transmit sequencer of the transmit engine, a signalencoded by the encoder; halting the sending step at an occurrence ofwhen the transmit sequencer sends a predetermined number of symbols thatare samples converted from the signal; receiving from the device andbatch processing, by a receive sequencer of the receive engine, thesignal decoded by the decoder; and halting the receiving step at anotheroccurrence of when the receive sequencer receives the predeterminednumber of said symbols.
 2. The modem according to claim 1 wherein saidinstructions further comprise: sequentially alternating betweenexecuting the sending and receiving steps.
 3. The modem according toclaim 1 wherein said instructions further comprise: simultaneouslyexecuting in parallel the sending and receiving steps.
 4. The modemaccording to claim 1 wherein said instructions further comprise:executing the sending and receiving steps at a symbol rate which isbased on a speed of the symbols arriving from the device.
 5. The modemaccording to claim 1 wherein said instructions carry out transmitoperations and receive operations.
 6. The modem according to claim 5wherein said instructions for carrying out the transmit and receiveoperations are synchronized by state indications in a shared data area.7. The modem according to claim 1 wherein said processor is a digitalsignal processor.
 8. The modem according to claim 1 wherein saidprocessor is a host processor.
 9. A method of operating a modem,comprising: storing in a memory system of a processor instructions foran encoder, a transmit engine, a decoder, and a receive engine, whereinsaid instructions comprise: sending to a device to which the modem is tocommunicate and batch processing, by a transmit sequencer of thetransmit engine, a signal encoded by the encoder; halting the sendingstep at an occurrence of when the transmit sequencer sends apredetermined number of symbols that are samples converted from thesignal; receiving from the device and batch processing, by a receivesequencer of the receive engine, the signal decoded by the decoder; andhalting the receiving step at another occurrence of when the receivesequencer receives the predetermined number of said symbols; andaccessing from the memory system and executing, by the processor, saidinstructions for processing said signal.
 10. The method according toclaim 9 wherein said instructions further comprise: sequentiallyalternating between executing the sending and receiving steps.
 11. Themethod according to claim 9 wherein said instructions further comprise:simultaneously executing in parallel the sending and receiving steps.12. The method according to claim 9 wherein said instructions furthercomprise: executing the sending and receiving steps at a symbol ratewhich is based on a speed of the symbols arriving from the device. 13.The method according to claim 9 wherein said instructions furthercomprise: carrying out transmit operations and receive operations. 14.The method according to claim 13 wherein said instructions for carryingout transmit operations and receive operations further comprise:synchronizing the transmit operations and the receive operations bystate indications in a shared data area.
 15. The method according toclaim 9 wherein the storing step further comprises the step of: storingsaid instructions in a memory system of a digital signal processor. 16.The method according to claim 9 wherein the storing step furthercomprises the step of: storing said instructions in a memory system of ahost processor.